In an analog TV (ATV) receiver, the automatic gain control (AGC) function is used to regulate the amplitude of the video signal and reject amplitude disturbances that occur in the transmission channel. Disturbances in the channel caused by constructive/destructive interference due to multipath propagation are often referred to as “flutter.” A common cause of flutter is moving objects, such as airplanes, that introduce a time varying flutter with frequencies ranging from a few Hz to a few kHz. Other causes of amplitude disturbances in the transmission channel are nonlinear distortions that convert frequency or phase of the signal into amplitude modulations and power supply modulations. These amplitude disturbances are commonly caused by poor TV signal amplifiers.
Many ATV signals have a periodic component that is used for synchronization of the video information to the display apparatus. Video information is displayed by drawing horizontal lines on the screen from left to right, and top to bottom. Each video line has a horizontal synchronization pulse (Hsync), and each video field (a multiple of video lines) has a vertical synchronization pulse (Vsync). The Hsync pulses have consistent amplitudes and are often used as reference levels for video AGCs. The video AGC measures the Hsync level, while ignoring the video content.
Because a single measurement is made for each line, the video AGC can be considered to be a discrete-time system, with a sample rate equal to the line frequency (typically 15,625 Hz or 15,750 Hz). The Hsync measurement is compared against a “target” that represents the desired amplitude level. The error between the Hsync measurement and the target is filtered and used to adjust the signal gain in an effort to drive the amplitude error to zero. Both feedback and feed-forward gain control can be provided.
FIG. 1 (Prior Art) is a block diagram of an embodiment 100 for AGC circuitry associated with an integrated TV receiver circuitry that demodulates incoming TV signals. The AGC circuitry includes a programmable feedback (FB) amplifier 102, a programmable feed-forward (FF) amplifier 106, and amplifier control circuitry 108. In operation, baseband TV signals 101 are provided to programmable FB amplifier 102, which outputs a signal to demodulator (DEMOD) 104. The demodulator (DEMOD) 104 outputs demodulated TV signals to the programmable FF amplifier 106, and the programmable FF amplifier 106 outputs demodulated and gain adjusted signals 107 that can be further processed, as desired. It is noted that the demodulator (DEMOD) 104 can be an analog TV (ATV) demodulator that removes analog signal modulations included on over-air television broadcasts.
The amplifier control circuitry 108 receives an output 109 from the demodulator (DEMOD) 104 at a detector 110. The detector 110 outputs an indication of the amplitude for the demodulated signal to a block 112 that converts the amplitude into a logarithmic value. For the feedback (FB) loop, this logarithmic value 113 is provided as a negative input to adder 124, which also receives a FB target value 126. The adder outputs a feedback error value 127 to a feedback filter (HFB) 128, which in turn provides a filtered feedback error value to an anti-log block 130. The anti-log block 130 converts the error value from a logarithmic value to a linear value, which is provided to ramp generator 132. Ramp generator 132 then provides a feedback amplitude correction signal 133 to the FB amplifier 102. For the feed-forward (FF) loop, the logarithmic value 113 is provided as a negative input to adder 114, which also receives a FF target value 116. The adder outputs a feed-forward error value 117 to a feed-forward filter (HFF) 118, which in turn provides a filtered feed-forward error value to an anti-log block 120. The anti-log block 120 converts the error value from a logarithmic value to a linear value, which is provided to ramp generator 122. Ramp generator 122 then provides a feed-forward amplitude correction signal 123 to the FF amplifier 106.
One aspect of the AGC circuitry is that the gain adjustment is made after the Hsync level measurement. This operation has the consequence of introducing a one-line delay in the AGC loop. This is represented in FIG. 2 (Prior Art).
FIG. 2 (Prior Art) is a signal flow diagram 200 that represents the transfer function for the demodulator AGC circuitry of embodiment 100 in the logarithmic domain. Delay block (Z−1) 204 represents a one-line delay caused by the discrete-time sampling of the Hsync level. The feedback loop filter response (HFB) is represented by block 228, and the feed-forward loop filter response (HFF) is represented by block 218. The adders 202 and 206 represent the programmable FB amplifier 102 and the programmable FF amplifier 106, respectively.
It is noted that the transfer function of the AGC loop is depicted in FIG. 2 (Prior Art) can be represented as follows:
      H    ⁡          (      z      )        =            1      -                        (                      H            ff                    )                ⁢                  z                      -            1                                      1      -                        (                      H            fb                    )                ⁢                  z                      -            1                              Each filter (HFB and HFF) can be considered a second order bi-quad. With the sampling delay in the loop, the general form of the overall AGC transfer function can be represented as follows:
      H    ⁡          (      z      )        =            1      +                        (                      b            1                    )                ⁢                  z                      -            1                              +                        (                      b            2                    )                ⁢                  z                      -            2                              +                        (                      b            3                    )                ⁢                  z                      -            3                              +                        (                      b            4                    )                ⁢                  z                      -            4                                      1      +                        (                      a            1                    )                ⁢                  z                      -            1                              +                        (                      a            2                    )                ⁢                  z                      -            2                              +                        (                      a            3                    )                ⁢                  z                      -            3                              +                        (                      a            4                    )                ⁢                  z                      -            4                              
A consequence of the sampling delay is that there is a non-zero amplitude response for some frequencies. In fact, some frequencies will have gain larger than 0 dB which means that the amplitude disturbance is increased at those frequencies. An example amplitude response is shown with respect to FIG. 4 (Prior Art), and this amplitude response represents the relative attenuation versus frequency that is seen by the amplitude disturbance
FIG. 3 (Prior Art) is a representative signal diagram 300 for an incoming TV signal in the form of a CVBS (Color, Video, Blanking, and Sync) signal including two horizontal lines 302 and 304. According to the standard CVBS signal format, a first horizontal line (N) 302 includes a horizontal sync tip signal 306, a color calibration signal (chroma burst) 308, and visible color content information 310. Similarly, the next horizontal line (N+1) 304 includes a horizontal sync tip signal 316, a color calibration signal 318, and visible color content information 320. The horizontal sync tip signals 306 and 316 are typically implemented as a negative pulse with a standard magnitude. As such, the magnitude of these horizontal sync tip signals 306 and 316 can be detected, as represented by blocks 312 and 322, and then can be used to determine a magnitude adjustment for the incoming CVBS signals. As depicted, this gain adjustment can be applied during remaining portion of the horizontal lines, as represented by elements 314 and 324. As such, regardless of how the gain is applied during the horizontal line (e.g., a step, a ramp, etc.), the effect of the gain change 314 made in the first horizontal line (N) 302 is not detected, as indicated by block 322, until the next horizontal sync tip 316. Similarly, the effect of the gain change 322 made in the next horizontal line (N+1) 304 would not detected until the following horizontal sync tip. This delay in applying the gain corrections can generate undesirable results in the resulting image.
FIG. 4 (Prior Art) is a signal diagram 400 for a frequency response 402 for a typical AGC in demodulation circuitry for TV signals, such as CVBS signals. The frequency response 402 includes a low-frequency notch region that blocks low-frequency signals and a high-frequency pass region that allows high-frequency signals to pass. As depicted, region 406 indicates a large negative response with respect to 0 db, which is represented by dashed line 420, and this region applies to low-frequency signals. In contrast, region 404 represents a smaller positive response with respect to 0 db that is applied to high-frequency signals. It is noted that the x-axis 410 represents frequency, and the y-axis 412 represents gain adjustments provided by the filter response 402. It is also noted that the cross-over frequency 422 can be 1 kHz, for example, although other cross-over frequencies could also be utilized.
The gain reduction provided by the low-frequency notch, as represented by region 406, operates to block low-frequency noise such as noise from power supplies and/or airplane flutter. The positive gain provided by the high-pass response, as represented by region 404, amplifies high-frequency noise. Traditionally, if the gain of the high-pass portion 404 is made smaller, then the gain reduction of the low-frequency notch portion 406 is undesirably narrowed, thereby increasing low-frequency noise and causing significant noise problems. In contrast, if the gain reduction of the low-frequency notch portion 406 is enhanced, then the high-pass portion 404 will become greater and further amplify the high-frequency noise, potentially giving rise to performance problems from high-frequency noise. Thus, there is a trade-off between the stop-band depth and/or width of the low-frequency notch and the high-frequency gain. As this depth and/or width of the notch is increased in region 406, the high-frequency gain in region 404 also increases. This increase can cause problems for downstream signal processing, such as video decoding, that rely on accurate synchronization to horizontal and vertical sync signals. While it is desirable to increase the depth and width of the low frequency rejection in region 406, such an adjustment is limited by the maximum high frequency gain in region 404 that can be tolerated by the downstream signal processing.